High-frequency tube apparatus with output direct - coupled - resonator filter



y 7, 1969 c; E. ROMIGUIERE 3,447,019

HIGH-FREQUENCY TUBE APPARATUS WITH OUTPUT DIRECT-COUPLED-RESONATOR FILTER I Filed Jan. 24, 1966 Sheet of 3 FIG.1

21. M 23 MR-2 Fl G, I m"om r z; y :2: f

p, L r, 1 5 from generator F T T to Load lw h k l l Transmitted on 12 2; 3

Power 100v. FIG .5

-1dB=8 O'l- (catcher c. vlty clone I Normal i zed Frequency 1 2.4 2.96 2h in 2n May 27, 1969 c. E. ROMIGUIERE HIGH-FREQUENCY TUBE APPARATUS WITH OUTPUT DIRECT-COUPLED-RBSONATOR FILTER Filed Jan. 24, 1966 mdi NoE

May 27, 1969 c. E. ROMIGUIERE 3,447,019

HIGH-FREQUENCY TUBE APPARATUS WITH OUTPUT Filed Jan. 24. 1966 DIRECT-COUPLED-RESONATOR FILTER Sheet 3 of 3 3,447,019 HIGH-FREQUENCY TUBE APPARATUS WITH OUTPUT DIRECT COUPLED RESONATOR FILTER Claude E. Romiguire, Montrouge, France, assignor to Societe Thomson-Varian, Paris, France, a company of France Filed Jan. 24, 1966, Ser. No. 522,565 Claims priority, application France, Jan. 25, 1965, 3 115 Int. (:1. iron 25/20 U.s. cl. 315-543 2 Claims ABSTRACT OF THE DISCLOSURE The present invention relates to microwave discharge tubes of the velocity-modulation type, particularly of the klystron type, having a high power and more particularly to improvements to the output circuit of these tubes, with the object of extracting a maximum power with a bandwidth wider than that obtained by the tubes of the prior art.

'It is known to extract the microwave electromagnetic energy from a velocity-modulation tube of the klystron type by coupling the output catcher cavity of the klystron tube directly to an output line, waveguide or coaxial line, by means either of a probe, or of a loop, or preferably by means of an inductive iris in the case of high-power tubes.

This kind of output circuit has a certain number of disadvantages due in'particular to the double requirements which the coupling between the output cavity and the output line has to meet.

Actually, the coupling of the output line to the corresponding cavity resonator defines the external Q (i.e. loaded Q) of the output cavity.

On the one hand, therefore, it determines directly the impedance olfered by the catcher cavity to the beam, and it is known that there is an optimum value of this impedance which enables a maximum electromagnetic energy to be abstracted from the beam, and hence a maximum efiiciency to be achieved. On the other hand, it is obvious that the external Q of the output cavity directly influences the bandwidth of the circuit.

When an engineer wishes to design a klystron with a wide bandwidth, he is therefore faced by the folowing dilemma: either to obtain the desired band by the adoption of a relatively low external Q, but at the same time to sacrifice efiiciency, or to retain a satisfactory efiiciency but at the expense of the bandwidth.

For very high-power amplifying klystrons, for example, it is common for the search for maximum efliciency to lead to the adoption of an external Q of the output catcher cavity resonator of the order of 25 to 30'. Now these values lead to relative bandwidths, measured at -1 db, of 3 to 4% at the most. The position is even worse for lower power tubes. For these latter tubes, the optimum external Q varies, from the point of view of chiciency, with the static impedance of the beam, which in- United States Patent creases when the power diminishes, for a given perveance of the electron gun.

On the other hand, in the majority of high power klystron tubes, the output cavity resonator is fixed, that is to say it is devoid of any tuning means, because it is difficult to realize a tuning device which will with-stand the very heavy currents which prevail in this cavity resonator. It is therefore impossible or at least not an advantage to increase the bandwidth of the tube by modifying the tuning of the output cavity by means of a tuning device.

The object of the invention is to increase the bandwidth of velocity-modulation tubes having a fixed output resonating cavity, solely by modifying the output circuit of the tube without changing its other members, and in particular without changing the interaction gap between the beam and the output catcher cavity resonator.

Prior attempts have been made to provide lklystron tubes having a high power output over a wide band of frequencies by coupling to the output catcher cavity an additional cavity resonator coupled itself to the output waveguide of the klystron tube, said additional cavity resonator being loaded to a Q value which is about .5 to .8 times the optimum Q value. An increase of bandwidth of the order of two times that possible with a single interacting cavity is obtatined from this embodiment.

In the present invention a direct-coupled-resonator filter is inserted between the output catcher cavity of the klystron tube and the output waveguide, this filter being an equal ripple filter having Chebyshev characteristics. This direct-coupled-resonator filter may comprise a single cavity resonator and, in this case, as will be seen, its structure and characteristics are quite different from the prior art additional cavity resonator and the bandwidth relative increase is substantially greater.

[In order to increase the bandwidth of a klystron tube having a fixed output resonant cavity, a plurality of resonant cavities directly coupled to one another are interposed between said output resonant cavity and the output waveguide. All said cavities, including the first coupled to the beam and the last coupled to the output line, resonate at the same frequency when they are considered individually, each then being assumed to be coupled to matched loads. The number of additional cavities depends on the desired increase in the bandwidth. The coupling coeflicient of the first additional cavity to the output catcher cavity of the klystron and the cOupling coefiicients of the additional cavities between themselves (the standardized reactances of the coupling irises in the case of coupling by means of irises) will be determined hereinafter.

The invention will now be described in detail with reference to the accompanying drawings in which:

FIGURE 1 is an equivalent diagram of a klystron intended for the theoretical explanation of the invention;

FIGURES 2a, 2b, 2c illustrate a novel arrangement of the output resonant cavities for a klystron according to the invention, the complete klystron being obtained by setting FIGURES 2a, 2b and 20 end to end as indicated by the arrows;

FIGURE 3- illustrates a section along the diametral plane passing through the axis of the klystron and the axis of the output waveguide of an output arrangement having two additional cavities; and

FIGURES 4 and 5 are diagrams explaining the determination of a prototype low-pass filter with an equal ripple and lumped constants and the manner of deriving therefrom a microwave filter having resonant cavities directly coupled representing the output arrangement for a klystron in accordance with the invention.

FIGURE 1 illustrates a generator G consisting of the portion of a klystron extending from the cathode to the interaction gap g of the output catcher cavity resonator and a load L consisting of the whole of the output catcher cavity resonator, the output waveguide and load proper. This figure enables the problem of the exchanges of energy between the generator G and the load L at the level of the gap to be studied.

If a conventional klystron tube is considered, in which the load is clearly determined and invariable and the drive frequency is caused to vary, the maximum power is obtained for a frequency i substantially equal to the cold resonant frequency of the output cavity resonator. Therefore the beam has not modified the resonant frequency of the output resonant cavity and it may be concluded from this that the internal impedance of the generator G in FIGURE 1 has no reactive component, that is to say it is purely real.

Taking the same klystron, it may be driven at the resonant frequency f of the output cavity; a variable impedance, the phase of which is caused to vary, is placed in the output waveguide and the output power is measrued. A succession of power maxima and minima are obtained as a function of the phase of the impedance. If the reflection coefficient of the impedance is caused to vary for the phase corresponding to the maxima, a maxi mum maximorum is obtained for a certain value of this reflection coeflicient. The fact that the maxima always take place for one and the same value of the phase of the impedance whatever be the reflection coefficient shows that the impedance of the load L is purely real at the frequency f The experiment of determining the maximum maximorum of power as a function of the phase and of the reflection coeflicient is repeated for different drive frequencies over a large range, adjusting the phase of the variable impedance and the reflection coeflicient each time. No appreciable variation in the output power is observed over the whole frequency band. It is concluded from this, that the generator G is aperiodic in this band.

The external Q of the output cavity resonator (sometimes called total effective Q) defined as the product times 211- of the ratio of the energy stored per cycle in the cavity resonator to the energy lost per cycle as a result of the coupling to the load, will be designated by Q It is known that the factor Q is such in which Q, is the Q resulting from the load due to the beam and Q is the normal Q in the absence of a load.

Just as there is an optimum phase for an impedance inserted in the output line, so there is an optimum Q of the output resonant cavity which will be designated by Q* which renders maximum the output power of the klystron at the resonant frequency of said output cavity resonator. It follows that there is an optimum load R* such that R*/Q* is equal to the quotient of the shunt resistance R by the unloaded Q of the cavity.

The existence of the optimum external Q* also follows from the following experiment. If the coupling of the output cavity to the load is too strong (Q Q"==) the very high frequency current of the beam reaches its saturation limit well before the voltage developed at the terminals of the gap of the output cavity resonator is at a maximum; thus, in the load L of FIGURE 1, a high current is obtained which depends little on the load but at a relatively low voltage, that is to say a reduced useful power. On the other hand, if the coupling is too weak (Q Q*) the very high frequency current of the beam develops a high retarding voltage in the output cavity resonator, which voltage rapidly reaches a critical value beyond which electrons are returned towards the cathode, which leads to disturbances in operation well before the current is completely modulated. A voltage is obtained at the terminals of the load L which is substantially independent of the load but with a relatively low current, hence a reduced useful power. The optimum coupling is a compromise selected in such a manner as to attain simultaneously the voltage saturation and the current saturation at the terminals of the gap.

At this point, the general result of the explanations which have preceded is that the generator G in FIGURE 1, constituded by the part of the klystron tube prior to the gap, may be represented by an equivalent constant current generator 21 in parallel with a resistor 22 of value R* or by an equivalent constant voltage generator 23 in series with a resistor 24 of value R These two equivalent generators are illustrated inside the rectangle G in FIGURE 1.

The load may be represented by a four-terminal network 25 which is purely passive and reactive, closed at the left through the resistor 22 of value R* and at the right through an impedance Z equal to the characteristic impedance of the output waveguide. The voltage standing wave ratio in the output waveguide which is assumed to be closed at the input through R*, should be substantially qual to unity throughout the whole pass-band desired for the tube.

The output resonant cavity having a Q when loaded by a resistor R' at the terminals of its gap may be replaced by a two-terminal network comprising a transmission cavity 26 (coupled at its two ends) coupled on the one hand to a matched load 27 of value R* and on the other hand to a filter 28 rendering the standing wave ratio in the output waveguide substantially equal to unity throughout the whole desired band.

If the standing wave ratio is equal to unity, the relative power transmitted is equal to unity. If is different from unity, the relative power transmitted is 4p*/ +P*) According to the invention, the filter 28 in FIGURE 1 is a filter with directly coupled resonant cavities having Chebyshev characteristics. It is derived from a Chebyshev prototype filter. The number of cavities in the filter depends on the desired increase in pass-band. The relative pass-band increases asymptotically with the number of cavities up to the asympotic limit of 3.91. The following table gives the number of cavities in the filter and the corresponding increase in relative pass-band.

Increase in relative pass-band Number of cavities The existence of an asymptotic limit will be proved in the following.

CALCULATION OF A NORMALIZED LOW-PASS CHEBYSHEV PROTOTYPE FILTER The Chebyshev prototype filter must have a number n of reactive elements equal to the number of the cavities of the direct-coupled-resonator filter. This point will be proved in the following.

The values of the reactances and the value of the output resistance of a normalized ladder network with Chebyshev characteristics having an equal ripple attenuation between 0 and a cut-off frequency /21r are given in paragraph 13.5 of the work: Network Analysis and Synthesis by Louis Weinberg, McGraw-Hill Book Company, Inc., New York, 1962. The first reactance designated by C in Weinbergs work and by g in the present specification is given by (formula 13.32 of Weinberg):

p a-ea where The cut-off frequency of a normalized Chebyshev filter is higher, .the larger are the reactances of the filter. Thus, for having the broadest bandwidth, one has to maximize these reactances, for example the value g of the first reactance.

CASE OF TWO ADDITIONAL CAVITIES (CURVE 103 OF FIG. 5)

Let us take the maximum attenuation a equal to -1 db which gives a level of maximum power efiiciency of 0.8. The total number of cavities is n=3 one catcher cavity plus two additional cavities); it results that the number of reactances of the Chebyshev prototype filter is n=3. They are designated by g g g in FIG. 4. It is known that if n is even, the last reactance before the load in the prototype filter is an inductance in series and if n is odd, this last reactance is a capacitor in parallel. In the present case the prototype filter thus comprises a capacitor g of capacity g farads, an inductance g of inductance value g henrys and a capacitor g of capacity g 'farads.

Let us take as varying parameter the quantity (ea-(e2) and 6 This table shows that the denominator of g, passes through a minimum equal to 0.6755 for p=33. The corresponding maximum for g is The maximal values of the other reactances g and g which all comprise the same denominator of a function of the denominator of g are correlatively determined. The bandwidth increase is /21r is the bandwith at -1 db of the catcher cavity resonator alone having a Q equal to Q*. Its attenuation bandwidth relationship is represented by curve of FIG. 5.

CASE OF ONE ADDITIONAL'CAVITY (CURVE 102 OF FIG. 5)

In this case the number of reactances of the Chebyshev prototype filter is n=2. They are designated by g and g A new table can be drawn up which is not reproduced here. The minimium of and the maximum of g 4 sin (qr/4) 1.168

The bandwidth increase is 2.42.

In the case of n==2, the calculation becomes simplified and can be made directly. In this case the following relationship exists between g and the normalized load resistance r (ratio of the load resistance to the generator resistance):

CASE OF ZERO ADDITIONAL CAVITY (CURVE 101 OF FIG. 5)

It is to be noticed that even in this case a bandwidth increase is obtained. It is not due to the cavities of the direct-coupled-resonator filter since there is no additional cavity, but to the design of the susceptance of the iris between the catcher cavity resonator and the waveguide which causes the catcher cavity resonator to act as an equal ripple cavity.

In this case the number of reactances of the Chebyshev prototype filter is n=1. This normalized filter comprises a generator resistance equal to unity, a shunt capacitor equal to unity and a load resistance equal to The bandwidth increase is 1.12.

DERIVATION OF A DIRECT-COUPLED-RlESONA- TOR FILTER FROM A NORMAL'IZED CHEBY- SHEV PROTOTYPE FILTER Determination of the normalized Chebyshev prototype filter The reactances of the normalized Chebyshev filter are determined as explained in the foregoing. The results are the following:

Case of three reactances 82:0.57 g '=1.93 r=0.44 Case of two reactances 81:2.42 g =0.35 r=0.38 Case of one reactance g =l.12 r=0.67 and all the normalized prototype filters have as their cut-off frequency /21r.

Determination of the Chebyshev prototype filter with a first reactance equal to unity In order to make easier the transformation of the prototype filter into a microwave waveguide filter, the reactances are divided by g and correlatively the frequencies are multiplied by g The reactances become:

Case of three reactances Case of two reactances r=0.3-8 Case of one reactance Determniation of the microwave filter by Cohns formulae g =L/r it n is even and g =Lr if n is odd ''1' 1 2 ('6 R f-X where w' =21rf' is the angular cut-off frequency of the passband of the prototype low-pass filter and k i are the limit wavelengths of the pass-band of the inductiveiris-coupled waveguide filter.

Derivation of a direct-coupled-resonator filter having two additional cavities The Cohns formulae give:

=4.17 (Susceptanco of the first iris) Where A is the wavelength in the Waveguide of the normalized type WR 284 for 3000 mHz.

If the catcher cavity resonator of the klystron tube without the output microwave filter resonates at a frequency equal to 3000 mHz. and has an optimum Q, Q*=5-0, the catcher cavity of the klystron tube provided with a three-cavity micro-wave filter has a Q of substantially 200, thus equal to approximately 4Q*, the first adcal ditional cavity resonator of the filter has a Q of substantially 2.60Q* and the second additional cavity resonator of the filter has a Q of substantially 0.28Q*.

Derivation of a direct-coupled-resonator filter having one additional cavity The Cohns formulae give:

If the catcher cavity resonator of the klystron tube without the output microwave filter resonates on a frequency equal to 3000 mHz. and has an optimum Q, Q*==50, the catcher cavity of the klystron tube provided with a twoeavity microwave filter has a Q of substantially 2 00, thus equal to approximately 4Q* and the additional cavity resonator of the filter has a Q of substantially 0.45Q*.

LIMIT OF THE BANDWIDTH INCREASE IN FUNC- TION OF THE NUMBER OF CAVITY RESONA- TORS OF THE MICROWAVE FILTER It is known (cf. H. W. Bode, Network Analysis and Feedback Amplifier Design, Van Nostrand, New York, 1959, page 3 66, formula 16.13) that, for a ladder filter of the type concerned where f designates the frequency.

R the resistance of the current source and g the value of the first reactance in the network of FIG. 4. Since R1=1 and g =1 co 1 I j c. \1

The broadest bandwidth would be obtained if a were constant and equal to a between- 0 and the cut-off frequency f' and infinite outside this range. Thus:

1 ii u-am whence O r =f 0 dr s 1 0 1 amp:

and

rnsx If a power efficiency of 0.8 is desired.

1 1 loge 5 1.61

For e- -==0.8 the cut-off frequency of the filter represented by curve of FIG. 4 is /21r. Thus the bandwidth increase limit is The bandwidth increase limit depends upon u according to the following table:

The bandwidth increase for e- =O.8 is consequently a minimum maximorum.

Referring now to FIGURE 2, the amplifying klystron comprises a tubular envelope 1 inside which is an appropriate high vacuum, for example of 10- mm. of mercury, created by means of an attached pump 2, for example an ionic pump, in gaseous communication with the space inside the envelope, by means of a suitable tubing 3.

An electron gun 4, arranged at one end of the envelope, serves to form and project 'an electron beam over a predetermined path, directed along the longitudinal axis of the envelope 1. An electron collector 5, arranged at the other end of the envelope, is adapted to collect the beam. A cooling fluid, for example water, is caused to circulate in suitable conduits (not illustrated) forming part of the structure of the electron collector 5.

A plurality of resonant cavities arranged along the envelope constitute the interaction structure. The input electromagnetic energy to be amplified is applied to the input resonator 7 through an input loop 9 and a coaxial line 11. The amplified output electromagnetic energy is extracted from the beam through an output cavity resonator 8 and transmitted to a suitable load (not illustrated) through an output iris 12, a first cavity resonator 13 a second iris 14 a second cavity resonator 13 and a third iris 414 and a waveguide 15. The guide 15 leads to a vacuum tight window 16 which is permeable to waves.

A solenoid (not illustrated) having the same geometrical axis as the envelope E1, surrounds this and generates a focussing magnetic field having an axial direction, serving to concentrate the beam along its axial path.

FIGURE 3 illustrates, on Ia larger scale, all the output cavity resonators. It shows the interaction gap 17 of the output catcher cavity 8 of the klystron and the projections 18 and 19 of the drift tubes. This cavity is coupled, through the iris 12, the normalized reactance of which is X 2 in the case of n=3, to .a first additional cavity 13 resonating at the same frequency f as the catcher cavity 8 and the electrical length of which is then through the iris 14 the normalized reactance of which is X to a second additional cavity 13 resonating at f and the electric length of which is and finally through the iris 14 the normalized reactance of which is X 4 to the output Waveguide 15.

The number of additional cavities may be diflerent from 1 or 2 and is given as a function of the increase in relative passband by the preceding table. The normalized reactances of the irises for coupling the cavities to one another are X X X etc. The length of the cavities is approximately equal to one half Wavelength less a small quantity depending on the normalized impedances of the coupling irises at the ends of each cavity.

Since many changes could be made in the above construction and many apparently widely different embodiments of the invention could be made without departing from the scope thereof, it is intended that all matter contained in the above description or shown in the accompanying drawings, shall be interpreted as illustrative and not in a limiting sense.

What I claim is:

1. A broadband output circuit for a high-power, velocity-modulated beam, amplifier tube comprising a catcher cavity resonator having an electron beam interaction region therein and a loaded quality factor substantially equal to four times the value Q* of said catcher cavity resonator required for maximum power transfer to said cavity resonator when directly coupled to a load two additional cavity resonators forming with said catcher cavity resonator an equal ripple microwave filter, the said catcher and additional cavity resonators being tuned to substantially the same frequency near the center of the desired pass-band for said tube, a first iris having a =normalized reactance of substantially 1/ 8.38 for coupling the first additional cavity resonator of said equal ripple filter to said catcher cavity resonator, a second iris having a normalized reactance of substantially l/6.71 for coupling the first and the second additional cavity resonators therebetween, a third iris having a normalized reactance of substantially 1/1.93 for coupling the second addi- 'tional cavity resonator to the load, the loaded quality ity-modulated beam,

factors of said first and second cavity resonators being substantially equal to respectively 2.6Q* and 0 .28Q*.

2. A broadband output circuit for a high power, velocamplifier tube comprising a catcher cavity resonator having an electron beam interaction region therein and a loaded quality factor substantially equal to four times the value Q* of said catcher cavity resonator required for maximum lpower transfer to said cavity resonator when directly coupled to a load, one additional cavity resonator forming with said catcher cavity resonator an equal ripple microwave filter, the

said catcher and additional cvity resonators being tuned to substantially the same frequency near the center of the desired pass-band for said tube, a first iris having a normalized reactance of substantially 1/ 8.53 *for coupling the cavity resonator of said equal ripple filter to said catcher cavity resonator, a second iris having a normalized reactance of substantially 1/ 2.60 for coupling the second additional cavity resonator to the load, the loaded quality factor of said additional cavity resonator being substantially equal to 0.45Q*.

References Cited UNITED STATES PATENTS 3,028,519 4/1962 Jepsen et al 315---5.43 3,173,103 3/1965 Bean et al. 33184 3,336,496 8/1967 Blinn 315-39.53

HERMAN KARL SAALBACH, Primary Examiner. SAXFIELD CHATMON, 111., Assistant Examiner.

US. Cl. X.R. 

